Ultrasonic imaging aberration correction system and method

ABSTRACT

An ultrasonic imaging system including an aberration correction system uses a harmonic component of the fundamental transmitted frequency for imaging, or for aberration correction, or both. By properly selecting the frequency pass bands of filters used in the image signal path and in the aberration correction signal path operating advantages are provided. The aberration correction values may be calculated concurrently with image formation.

This application is a divisional of U.S. patent application Ser. No.09/383,518, filed Aug. 26, 1999, U.S. Pat. No. 6,131,458 which is adivisional of Ser. No. 09/061,082, filed Apr. 15, 1998, now U.S. Pat.No. 6,023,977, filed Apr. 15, 1998, which is a continuation-in-part ofU.S. patent application Ser. No. 08/904,859, filed Aug. 1, 1997(abandoned).

BACKGROUND OF THE INVENTION

This invention relates to ultrasonic imaging systems and methods whichutilize ultrasound echo information at a harmonic of the fundamentalfrequency of transmitted ultrasonic energy either for image formation orfor aberration correction value estimation.

In an ultrasound imaging system, the velocity of sound is usuallyassumed constant in tissue in order to calculate time delays in formingacoustic beams from transducer arrays. However, the velocity ofultrasound waves in body tissues varies over a wide range. Therefore,ultrasound waves experience wavefront distortion, which disruptsdiffraction patterns and produces image artifacts.

Several approaches have been proposed to correct for sound velocityinhomogeneities in tissue. One approach is to model the sound velocityinhomogeneities as a simple phase screen at or near the face of thetransducer. Under this condition, sound velocity inhomogeneities resultin time-of-flight errors (i.e., phase aberrations) and the receivedsignal in one channel can be approximated by a time-delayed replica ofthe signal received by another channel. Therefore, phase aberrations canbe estimated (1) by determining the peak position in thecross-correlation of signals received by two adjacent channels orsubarrays (S. W. Flax and M. O'Donnell, “Phase aberration correctionusing signals from point reflectors and diffuse scatterers: basicprinciples,” IEEE Trans. Ultrason., Ferroelect. Freq. Contr., vol. 35,no. 6, pp. 758-767, 1988), or (2) by maximizing speckle brightness viatime delay adjustment (L. F. Nock, G. E. Trahey, and S. W. Smith, “Phaseaberration correction in medical ultrasound using speckle brightness asa quality factor,” J. Acoust. Soc. Am., vol. 85, no. 5, pp. 1819-1833).Another proposed method estimates aberrating delays by utilizing arrayredundancy in spatial frequency (D. Rachlin, “Direct estimation ofaberrating delays in pulse-echo imaging systems,” J. Acoust. Soc. Am.vol. 88, no. 1, pp. 191-198, 1990).

The validity of the near field thin phase screen model has beenquestioned, based on the fact that waveform distortions in addition totime delay errors have been observed. (D. L. Liu and R. C. Waag,“Correction of ultrasonic wavefront distortion using backpropagation anda reference waveform method for time shift compensation,” J. Acoust.Soc, Am., vol. 96, no. 2, pp. 649-660, 1994). These waveform distortionshave been explained by modeling the acoustic velocity inhomogeneities asdistributed throughout the region between the transducer and the targetor by putting the phase screen at a distance away from the face of thetransducer. Various methods have been proposed to correct fordistributed aberrations (or displaced phase screens). They include aback propagation method (Liu, et al., supra), a total least squares(TLS) based approach called PARCA (S. Krishnan, P. C. Li, and M.O'Donnell, “Adaptive compensation of phase and magnitude aberrations,”IEEE Trans. Ultrason., Ferroelect. Freq. Contr., vol. 43, no. 1, pp.44-55, 1996), and a time reversal focusing technique (M. Fink, “Timereversal focusing in ultrasound: basic principles,” IEEE Trans.Ultrason., Ferroelect. Freq. Contr., vol. 39, no. 5, pp. 555-566, 1992).

Recently, other alternative approaches have also been developed tocorrect for distributed aberrations. They include a phase conjugationapproach (G. C. Ng, P. D. Freiburger, W. F. Walker, and G. E. Trahey, “Aspeckle target adaptive imaging technique in the presence of distributedaberrations,” IEEE Trans. Ultrason., Ferroelect. Freq. Contr., vol. 44,no. 1, pp. 140-151, 1997), which independently corrects for time delayerrors for each frequency component, and an inverse filtering approach(Q. Zhu and B. Steinberg, “Deaberration of incoherent wavefrontdistortion: an approach toward inverse filtering,” IEEE Trans.Ultrason., Ferroelect. Freq. Contr., vol. 44, no. 3, pp. 575-589, 1997),which compensates for both phase and amplitude distortion in thefrequency domain.

One major factor that determines the efficacy of all the methodsmentioned above is the quality of the transmit beam profile. A goodtransmit beam profile (narrow mainlobe and low sidelobes) improves boththe image quality and the estimation accuracy. It has been shown thatharmonically generated transmit beam profiles have lower sidelobes andare less sensitive to the phase aberrations that are present. (T.Christopher, “Finite amplitude distortion-based inhomogeneous pulse echoultrasonic imaging,” IEEE Trans. Ultrason., Ferroelect. Freq. Contr.,vol. 44, no. 1, pp. 125-139, 1997).

The above-referenced Christopher article speculates as to aberrationcorrection in a harmonic ultrasonic imaging system, but provides nodetails as to the structure or operation of any such system.

Wright U.S. Pat. No. 5,570,691, assigned to the assignee of the presentinvention, discloses one particularly advantageous aberration correctionvalue estimation system in which ultrasonic energy from a single firingor transmit event is used both in the formation of the ultrasonic imageand in the calculation of aberration correction values. In this way, theneed for separate aberration correction lines or frames can beeliminated.

Johnson U.S. Pat. No. 5,456,257 discloses systems for ultrasonicallydetecting contrast agents. In one disclosed embodiment signals fromcollapsing microbubbles included in a contrast agent are used tocalculate delay adjustments intended to correct for tissue aberration.Little detail is provided regarding the structure of the disclosedsystem, and contrast agent is essential for operation of the disclosedsystem.

SUMMARY OF THE INVENTION

The present invention is defined by the appended claims, and nothing inthis section should be taken as a limitation on those claims. By way ofgeneral introduction, the present invention relates to aberrationcorrection in ultrasonic imaging systems in which a harmonic of thefundamental transmitted frequency is used for imaging, or for aberrationcorrection, or both. As discussed below, by properly selecting thefrequency pass bands of filters used in the image signal path and in theaberration correction signal path, improved systems can be provided withsubstantial operating advantages. Certain of the embodiments describedbelow calculate aberration correction values concurrently with imageformation.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an ultrasound imaging system suitable foruse with this invention.

FIGS. 2-14 are block diagrams of alternative embodiments of the receivesignal processor of FIG. 1.

FIG. 15 is a graph showing three alternative beam profiles.

FIGS. 16 and 17 are graphs showing fundamental and harmonic componentsof received and filtered received waveforms, respectively.

FIGS. 18 and 20 are block diagrams of alternate embodiments of receivesignal processors suitable for use in the system of FIG. 1.

FIGS. 19 and 21 are flow charts illustrating operation of theembodiments of FIGS. 18 and 20, respectively.

DETAILED DESCRIPTION OF THE PRESENTLY PREFERRED EMBODIMENTS GeneralDiscussion

As discussed above, several prior-art techniques measure delaydifferences from transducer element to element (or subarray to subarray,or element to subarray) to obtain aberration correction estimates. Theseprior-art techniques depend on obtaining a receive signal of measurableamplitude across the face of the transducer array. Specifically, thereshould not be significant dropouts of signals due to destructiveinterference (caused by the aberrations) at adjacent transducer elementsor subarrays, or the resulting signal to noise ratio will suffer. Lowerfrequency signals will be less susceptible to such destructiveinterference, resulting in more uniform amplitudes at adjacent channelsat the face of the transducer.

In a typical aberration correction technique, the relative phasereceived at adjacent elements (or subarrays or subarray/element pairs)is measured and converted to a time delay. Using such a technique,aliasing will occur when the aberration causes a relative delay in thereturning wavefront greater than half a wavelength between adjacenttransducer elements or subarrays. Specifically, if the aberration causesmore than half a wavelength in delay, the aberration correction will beunderestimated. This will be less likely to occur at lower frequencies(longer wavelengths), because a greater time delay error can be measuredwithout aliasing using a lower frequency than using a higher frequency.

As a result, more accurate aberration correction estimates can often beobtained using relatively lower receive frequency information.

On the other hand, higher frequency imaging generally provides forbetter detail and contrast resolution, assuming that adequatepenetration can be maintained. As is well known, higher frequenciesattenuate in tissue more quickly than lower frequencies. Accordingly, inthe absence of aberrations, the use of higher frequencies and theresulting narrower beam profiles will generally result in a betterimage.

In view of the foregoing, one aspect of the present invention isdirected to taking advantage of different properties of differentfrequency bands or components of the ultrasonic echo information byusing different frequency bands for imaging and for aberrationcorrection to improve image quality.

As discussed in detail below, selected embodiments of the inventionachieve this objective using tissue harmonic imaging in combination withaberration correction techniques. In tissue harmonic imaging, theultrasound system images a harmonic response generated by the tissue inresponse to a transmitted fundamental ultrasound waveform from atransducer. This harmonic response increases as a function of theamplitude of the transmitted waveform. Accordingly, as the transmittedwaveform propagates toward the transmit focus, the amplitude of thewaveform increases, thus increasing the generation of the harmonicresponse of the tissue. This results in a generated harmonic frequencytransmit beam profile at the transmit focus that has preferredcharacteristics, and that is less influenced by tissue aberrations, thanthe transmit beam profile that would have been obtained had thetransducer transmitted a signal at the harmonic frequency. Generally, intissue harmonic imaging no additional non-linear contrast agent is addedto the tissue at any time during the ultrasound examination session.

This principle is schematically illustrated in FIG. 15, in which thetop, dotted line represents an ideal (absent aberration) transmit beamprofile resulting from transmission of a waveform centered at afundamental frequency, such as a center frequency of 2.5 MHz. Themiddle, dashed line represents an ideal transmit beam profile resultingfrom transmission of a waveform centered at a harmonic frequency such asa center frequency of 5 MHz. The lower, solid line represents an idealtransmit beam profile obtained from the harmonically generated frequencyband, i.e. transmission of ultrasonic energy at the fundamentalfrequency and reception of ultrasonic echo information at the harmonicfrequency. The solid line shows a profile having the desirablecharacteristics of a well-formed main lobe (which is narrower than thatof the top profile) in combination with low sidelobes.

When aberrations are present, both the fundamental transmit beam profile(from which the harmonic frequencies are generated), and the bodygenerated harmonic beam profile are less affected by aberrations than atransmit beam profile generated by the transducer at the harmonicfrequency.

Preferred embodiments of this invention use information from thefundamental and the body generated harmonic beam profiles to createimages and to estimate aberration correction values.

In order to obtain the benefit of the body-generated harmonic transmitbeam profile described above, it is important to suppress or excludetransmitted ultrasonic energy in the harmonic frequency band. This canbe done by shaping the transmit waveform and/or by filtering the outputof the transmitter before it is applied to the transducers. Methods andapparatus for achieving a well-formed shaped transmit beam are disclosedin U.S. patent applications Ser. Nos. 08/642,528, filed May 3, 1996, nowU.S. Pat. No. 5,740,128; Ser. No. 08/893,287, filed Jul. 15, 1997, nowU.S. Pat. No. 5,833,614; Ser. No. 08/893,271, filed Jul. 15, 1997, nowabandoned and Ser. No. 08/893,150, filed Jul. 15, 1997, now U.S. Pat.No. 5,913,823. Resulting transmit waveforms will be referred to hereinas “shaped fundamental” waveforms. In these shaped fundamentalwaveforms, the second harmonic component is suppressed by at least 20 dBthroughout a bandwidth of 10% around the second harmonic frequency(i.e., if the second harmonic is at 5 MHz, then from 4.75 to 5.25 MHzthe signal is suppressed by 20 dB or more with respect to the signallevel at the fundamental frequency). The second harmonic is preferablysuppressed by at least 35 dB throughout a bandwidth of 15% around theharmonic frequency, and more preferably is suppressed by at least 45 dBthroughout a bandwidth of 20% around the harmonic frequency. Mostpreferably, the second harmonic is suppressed by at least 55 dBthroughout a bandwidth of 25% around the harmonic frequency.

FIG. 1 shows a block diagram of an ultrasonic imaging system 10 that canbe used to implement this invention. The system 10 includes a transmitbeamformer 12 that applies transmit waveforms via a transmit/receiveswitch 14 to a transducer array 16. The transducer array 16 operates asa phased array, and the transmit beamformer 12 times the transmitwaveforms as appropriate to focus transmitted ultrasonic energy from thetransducer array 16 at a desired direction and range.

The transmit beamformer 12 can be implemented in any suitable mannerincluding either analog or digital beamformers. The presently preferredtransmit beamformer 12 takes the form described in U.S. patentapplication Ser. No. 08/673,410, filed Jul. 15, 1996, assigned to theassignee of the present invention. The transmit beamformer 12 can alsotake the form of any of the transmit beamformers described in theabove-identified U.S. patent applications Ser. Nos. 08/642,528,08/893,287, 08/893,271 and 08/893,150.

The transducer array 16 generates ultrasonic energy that is transmittedinto a target 18, typically a tissue of the subject. In alternativeembodiments, the transducer array 16 may be a one dimensional, one and ahalf dimensional or two dimensional array, as appropriate for theapplication. The target 18 may optionally include a non-linear contrastagent added to increase the harmonic response of the tissue. However, inmany applications it is preferable to avoid the use of added non-linearcontrast agent, and to rely on tissue harmonic imaging as describedabove, in which the body tissue itself generates ultrasonic energy at aharmonic of the fundamental frequency transmitted by the transducerarray 16 and no additional non-linear contrast agent is introduced intothe patient at any time during the ultrasound examination session. Suchtissue harmonic imaging can be used in all of the specific embodimentsdescribed below.

Ultrasonic echo information received by the transducer array 16 isrouted by the switch 14 to a receive signal processor 20. The receivesignal processor 20 typically includes a receive beamformer and filtersas described in detail below. The receive signal processor 20 appliesoutput signals to an adaptive focusing control system 22 and to an imageprocessor 24.

As shown in FIG. 1, the adaptive focusing control system 22 includes anaberration correction value estimator 26 and an adaptive focus processor28. The estimator 26 and the processor 28 preferably take the formdescribed in U.S. Pat. No. 5,570,691, assigned to the assignee of thepresent invention. Generally speaking, in this embodiment the aberrationcorrection value estimator 26 estimates aberration correction valuesboth in terms of focusing phase or delay and amplitude. The processor 28stores the aberration correction values determined by the estimator 26and uses these stored values to correct both focusing and amplitudeaberrations associated with both the transmitted and the receivedultrasonic echo information. In this embodiment, correction values forfocusing phase or delay are determined by cross-correlating signals fromadjacent regions of the transducer array 16.

The aberration correction value estimator 26 can use partial summationsignals from subarrays of transducer elements in calculating aberrationcorrection values. When such subarrays are used, the signals from two ormore adjacent transducer elements are summed to form subarray sums. Thesignals being summed have already been adjusted by the normal focusingdelays and apodization values by the receive beamformer and have alsobeen adjusted by previous aberration delay estimates and apodizationcorrection values from the apodization focus processor 28. (As usedherein “normal” as applied to focusing delays and apodization valuesmeans values not taking aberrations into account.) In the aberrationcorrection value estimator 26 the subarray sums are processed todetermine the residual error (whether from delay, apodization, or both)across all subarrays. While it is preferred to use subarrays to reducehardware complexity, in alternative embodiments the aberrationcorrection value estimator 26 may respond to individual transducerchannels rather than subarrays.

In addition to measuring phase error of the subarray sum signals, theaberration correction value estimator 26 also measures the amplitude ofthe signals in order to determine the amplitude variations betweenvarious elements or subarrays in the array. At one or more specificranges, the digital signal values are averaged to determine an averagesignal level over a few cycles. The magnitude of the signals is used inthis calculation to eliminate cancellation due to positive and negativephases.

In particular, the system 10 includes a beamformer controller 30 thatgenerates normal focusing data and apodization data as shown. The normalfocusing data is applied to a summer 32 that also receives focusingaberration correction values from the processor 28. The normal focusingdata supplied by the beamformer controller 30 is determined bygeometrical considerations assuming no tissue aberration. The output ofthe summer 32 combines normal focusing data appropriate for the selectedacoustic line as well as focusing aberration correction values asdetermined by the estimator 26 and the processor 28. The sum is appliedas corrected transmit focusing data to the transmit beamformer 12 and ascorrected receive focusing data to the receive signal processor 20.

Similarly, the system 10 includes a multiplier 34 that receives inputsignals both from the beamformer controller (normal apodization data)and from the processor 28 (amplitude aberration correction values). Theproduct of these two inputs is supplied as corrected transmitapodization data to the transmit beamformer 12 and as corrected receiveapodization data to the receive signal processor 20.

The aberration correction value estimator 26 may operate for all scanlines or for only a subset of scan lines and may operate at one range(typically the transmit focus) or multiple ranges. If aberrationcorrection values are derived from multiple lines and/or ranges, theresulting data is preferably stored in a table which is usedsubsequently to determine aberration correction values by means ofinterpolation for example. See for example, U.S. Pat. No. 5,570,691,assigned to the assignee of the present invention. Simpler systems inwhich aberration correction values are only calculated for as little asone range and one acoustic line are also allowed for in this invention.As part of its operation, the adaptive focus processor 28 mayinterpolate focusing and apodization correction values detected forspecific transducer subarrays to determine correction values to beapplied to individual array channels.

The aberration correction value estimator preferably responds to agating signal from the beamformer controller 30 such that the aberrationcorrection value estimator 26 and the image processor 24 are bothresponsive to ultrasonic echo information associated with the sametransmit event or firing of the transducer array 16. Preferably, theaberration correction value estimator 26 operates only on apredetermined range of the received ultrasonic echo information. This isaccomplished by a gating signal as described in U.S. Pat. No. 5,570,691.A suitable embodiment for the aberration correction value estimator isdescribed in U.S. Pat. No. 5,570,691 at columns 21-22. See theaberration value estimator G-502 of FIG. 6 of that patent. Gating of theestimator is shown in FIGS. 4 and 5 (see Reference Nos. 550 and 572,574, respectively). Other aberration value correction techniques may beused with this invention.

In the preferred embodiment of FIG. 1 the normal and corrected delay andapodization values are combined by the summer 32 and the multiplier 34prior to being sent to the transmit beamformer 12 and the receive signalprocessor 20. In alternative embodiments the combining can take place aspart of the beam formation process.

As explained above, the receive signal processor 20 also suppliesbeamformed signals to the image processor 24. The image processor 24 cantake any suitable form, and can be constructed as an analog, digital orhybrid system. The image processor 24 preferably includes brightness orintensity processing circuitry 36 and motion processing circuitry 38.The elements 36, 38 can take any suitable form, and if desired themotion processing circuitry 38 may be deleted. The processing circuitry36 can be implemented by any known or later developed type of imageprocessing that involves displaying the information with brightnessrepresenting intensity, such as known B-mode and/or M-mode ultrasoundimage processing. This type of image is typically but not alwaysdisplayed as a gray scale. The processing circuitry 38 may include anydesired type of motion processing such as color Doppler or time shiftinformation processing, and is typically but not always displayed incolor. Output signals from the elements 36, 38 are stored in a memory 40prior to scan conversion in a scan converter 42. The scan convertedoutput of the image processor 24 is applied to a conventional display44. The brightness processing circuitry 36 and the motion processingcircuitry may be used in combination for a single image.

As explained below, the receive signal processor 28 can be adapted tosupply the desired frequency components of the received signal to theaberration correction value estimator 26 and to the image processor 24.In one preferred embodiment, a shaped fundamental transmit waveform istransmitted (centered at the fundamental frequency f₀) and the returnsignal is selectively received at a harmonic frequency band (centered at2f₀) and used to form an ultrasound image. Ultrasonic echo informationmay also be optionally selectively received at the fundamental frequencyband (centered at f₀), and this information may also be optionally usedin the formation of the image. As discussed below, in certaincircumstances it may be desirable to combine the fundamental andharmonic components of the echo information in the image processor 24.

Similarly, in various embodiments, aberration correction values areestimated from information obtained from the fundamental frequencycomponent, the harmonic frequency component, or both. Depending upon theapplication, it may be preferable to allow great flexibility in thefiltering characteristics of the receive signal processor 20.Alternately, simpler but less flexible implementations are possible inwhich fixed filters are provided in the two signal paths provided by thereceive signal processor 20 for the adaptive focusing control system 22and the image processor 24. For example, in selected ones of thepreferred embodiments discussed below a harmonic filter is used in thesignal path for the image processor 24 and a fundamental filter is usedin the signal path for the aberration correction value estimator 26.

The aberration estimation can be determined using a transmit firingdedicated to the aberration estimate, and not used for creating theimage. Alternatively, a wide band signal can be received, and thebandwidth of the receive signal filtered appropriately for the image andaberration correction signal paths. For example, the entire bandwidth ofthe receive signal or the fundamental band of the receive signal can beused for image formation, while only the fundamental or harmoniccomponent of the receive signal can be used for aberration valueestimation.

Specific Embodiments

The receive signal processor 20 can take many forms, and the followingsections discuss selected embodiments. In FIGS. 2-14 the illustratedreceive signal processors have been designed to function within theultrasonic imaging system 10 of FIG. 1, receiving input signals from theswitch 14, the summer 32 and the multiplier 34, and providing outputsignals to the adaptive aberration value estimator 26 and the imageprocessor 24.

Turning now to FIG. 2, the receive signal processor 50 includes areceive beamformer 52. The receive beamformer 52 is preferably a digitalbeamformer, but it may be implemented as an analog beamformer ifdesired. Ultrasonic echo information from the transducer array 16 andthe switch 14 passes through high voltage clamping circuits for receiverprotection and preamplifiers. These preamplified signals are then passedthrough time varying amplifiers prior to conversion to digitalquantities in an analog to digital converter. The receive beamformer 52typically operates with base band signals in the form of in phase andquadrature (I, Q) quantities. The focusing delay profile is dynamicallyupdated as the focal depth evolves for signals arriving fromprogressively deeper regions of the target.

The receive beamformer 52 receives aberration corrected delays andapodization values, which may if desired change dynamically (i.e. delaysassociated with aberration correction which may change as a function ofelapsed time from the start of the transmit event are added todynamically changing focus delays). In this way, near ideal focal delaysare applied to obtain near perfect focusing, despite the presence ofaberrating tissue delays. Amplitude correction values are also appliedto the receive signals by means of scaling multipliers. Alternately,correcting amplitude factors may be applied by means of the controlledgain amplifiers, in which case the gain is determined by an appliedvoltage derived from the amplitude correction values.

In this embodiment the receive beamformer 52 is a multiple beambeamformer that simultaneously forms appropriately delayed outputs fortwo respective beams from a single transmit event. The first of thesebeams is demodulated with a frequency 2f₀ (schematically illustrated bythe filter 54) and then applied via a summer 56 which sums all channelsto a filter 58. In this case the filter 58 is a baseband filter thatpasses signals energy that was originally centered at the harmonicfrequency 2f₀ prior to demodulation. The beamformed receive signal inthe harmonic frequency band is applied as an input to the imageprocessor 24 of FIG. 1.

The second beam formed by the receive beamformer 52 is demodulated withthe fundamental frequency f₀, schematically indicated by the filter 60.The time delayed signals from two or more adjacent transducer elementsare summed in a partial summer 62 and applied as an input to a low passfilter 64 which passes signal energy that was originally centered at thefundamental frequency f₀ prior to demodulation. This demodulated,filtered, partially summed signal obtained from a receive signal at thefrequency band centered at the fundamental f₀ is applied as an input tothe aberration correction value estimator 26 of FIG. 1.

In the embodiment of FIG. 2 the two beams formed by the receivebeamformer 52 are focused using both time and phase adjustments. Theleast significant bits of the time delay are converted to phaseadjustments. While time adjustments are independent of frequency, phaseadjustments are not. Because the receive beamformer 52 produces twobeams, each can be optimally focused using phase adjustments asappropriate for the harmonic and the fundamental receive information,respectively.

The filters in the receive beamformer 52 can be regarded as coarseadjustments, and the filters 58, 64 may be regarded as fine adjustmentsto signal quality. If desired, the filter 54 or the filter 58 can bedeleted. Similarly, if desired, the filter 60 or the filter 64 can bedeleted. As explained above, if the aberration correction valueestimator is intended to function on individual transducer channels thepartial summer 62 can also be eliminated. In another embodiment relatedto that of FIG. 2, the aberration correction value estimator isresponsive to the entire unfiltered broadband receive signal, which ismade up primarily of ultrasonic echo information in the fundamental passband.

In receive signal processors such as the processor 50, a filter such asfilter 58 in the imaging path can be implemented as the baseband filter,such as the filter described in U.S. patent application Ser. No.08/434,160, filed May 2, 1995, and assigned to the assignee of thisinvention. Since the signal applied to the filter 58, 64 is downshiftedto baseband (i.e. DC) in the receive beamforming process, the fineadjustment filter 58, 64 is typically implemented as a low pass filterabout DC.

The embodiment of FIG. 2 is particularly advantageous in situationswhere aliasing or dropout problems are encountered in the eventaberration correction estimation is attempted at the harmonic. It isanticipated that the embodiment of FIG. 2 will be well suited for B-modeor M-mode imaging.

The embodiment of FIG. 2 is also anticipated to be well adapted forcalculating error correction values when aberration correction is neededin regions of relatively greater depths. Since lower frequencies areless attenuated by body tissues than higher frequencies, a better signalto noise ratio (SNR) will be obtained in such circumstances if thefundamental frequency component rather than the harmonic frequencycomponent is used to calculate aberration corrections. In particular,there is a significant difference in the SNR level in the correction andimaging paths due to the facts that SNR varies as the square root of thenumber of channels, and the number of channels is substantially smallerin the correction path than the imaging path. For this reason, thequality of the information applied to the aberration correction valueestimator is not as high as that applied to the image processor, and itis therefore particularly advantageous to use the fundamental componentof the ultrasonic echo information for aberration correction.

FIG. 3 shows a block diagram of a receive signal processor 70 that alsoapplies received ultrasonic echo information in the fundamental passband to the aberration correction value estimator, and receivedultrasonic echo information in the harmonic pass band to the imageprocessor. In this case the receive signal processor 70 includes areceive beamformer 72 that operates as described above except that itproduces a single output signal, a wide band signal including both thefundamental and harmonic frequency components. This wide band signal isoptionally partially summed to form a subarray summation signal which isapplied to an optional filter 74 designed to pass the fundamentalcomponent near f₀ and to block the harmonic component near 2f₀. Theoutput of the optional filter 74 is applied to the aberration correctionvalue estimator 26 of FIG. 1. The partial summation signals are appliedto a summer 76, and the output of the summer 76 is applied to a filter78 designed to pass the harmonic component near 2f₀ and to block thefundamental component near f₀. As explained above, the filter 74 may bedeleted if desired. As in the embodiment of FIG. 1, aberrationcorrections are estimated based on the fundamental component of theultrasonic echo information, while the image is formed based on theharmonic component of the ultrasonic echo information.

FIGS. 4 and 5 relate to receive signal processors 80, 90, respectively,which apply ultrasonic echo information from the harmonic frequency band(2f₀) to the aberration correction value estimator 26 of FIG. 1 andultrasonic echo information from the fundamental frequency band (f₀) tothe image processor 24 of FIG. 1. The receive signal processor 80 ofFIG. 4 includes a dual beam receive beamformer 82 that generates tworeceive beams in response to each transmit event. The first beam isdemodulated to emphasize ultrasound echo information near thefundamental frequency f₀ (schematically indicated by the filter 84), andthe second receive beam is demodulated to emphasize received ultrasonicecho information at the harmonic (schematically indicated by the filter86). The first receive beam demodulated at the fundamental frequency f₀is applied to a summer 88 and via a filter 90 that suppresses theharmonic component applied to the image processor. The second receivebeam is applied via an optional partial summer 92 and a filter 94 thatsuppresses the fundamental component applied to the aberrationcorrection value estimator. As before, one or both of the filters 86, 94may be deleted and one or both of the filters 84, 90 may be deleted.Also, the partial summer 92 is not required in all applications. Thereceive beamformer 82 of FIG. 4 provides many of the advantages of thedual beam receive beamformer 52 of FIG. 2.

The receive signal processor 100 of FIG. 5 uses a receive beamformer 102that generates a single output delayed and amplitude adjusted asdescribed above. An optional partial summer 103 is included, and theoutput of the partial summer 103 (if present) is supplied via a summer104 and a fundamental filter 105 as an input to the image processor.Partially summed signals from the partial summer 103 (if present) areapplied via a harmonic filter 106 as an input to the aberrationcorrection value estimator.

It is anticipated that the embodiments of FIGS. 4 and 5 will be wellsuited for color Doppler (F-mode) applications, where it is advantageousto use lower frequencies for imaging and higher frequencies forcalculating aberration corrections. The use of low frequencies forimaging in color Doppler applications is important because sensitivityis a greater issue. Received fundamental Doppler signals are at a muchlower signal level than received fundamental B-mode imaging signals,e.g. 40 dB lower at a given fundamental frequency. Because of greaterattenuation at higher frequencies, the lower frequencies will be oflarger amplitude and therefore easier to detect, providing betterinformation for imaging. On the other hand, harmonic signals used forcalculating aberration corrections, which are the sum of one or a smallnumber of channels, are about 40 dB below the B-mode imaging signallevel. Thus, harmonic correction signals are comparable to the signallevel for color Doppler imaging. If the SNR is adequate for colorprocessing the fundamental frequency component, it should also beadequate for aberration correcting the harmonic frequency component. Ifthe SNR is adequate, it is preferred to perform aberration correctioncalculations using the harmonic component, because the harmonicallygenerated transmit beam profile is well formed, and has lower sidelobes,giving a more accurate result.

The harmonic component is preferred for aberration correction incircumstances where aliasing is not a problem, and the signal to noiseratio of the harmonic beam is above a threshold value. Wherecircumstances permit, it is generally preferred to use the harmoniccomponent for aberration calculations.

The composite, or overall beam profile (sometimes referred as the pointspread function) determines the sensitivity of an ultrasound system. Thecomposite beam profile can be approximated by multiplying the transmitbeam profile and the receive beam profile. For each subarray of thetransducer array, the receive beam profile is rather unfocused, since itis created from a relatively small number of channels (1 to 4 in thepreferred embodiment, although a larger number of channels can be usedper subarray in alternative embodiments). On the other hand, a muchlarger number of elements are used on transmit, with the result that thetransmit beam profile is much more focused (localized). For this reason,when the transmit and receive beam profiles are multiplied together tocreate the composite profile of a subarray signal, the composite profilewill look very much like the transmit beam profile. As discussed abovein conjunction with FIG. 15, the harmonically generated transmit beamprofile is desirable to use whenever possible, because it has awell-formed main lobe and small side lobes. Accordingly, it should givemore accurate results in calculating aberration correction values,providing that there is no excessive aliasing and that the signal tonoise ratio is adequate, i.e. that the signal has not been overlydiminished due to attenuation.

FIGS. 6 and 7 relate to receive signal processors 110, 120,respectively, that apply harmonic signals to both the aberrationcorrection value estimator and the image processor. The receive signalprocessor 110 of FIG. 6 is similar to the processor 90 discussed above,except that both of the filters 112, 114 are harmonic filters that passreceived ultrasonic echo information at the harmonic rather than thefundamental (2f₀ rather than f₀ in this example). The receive signalprocessor 120 of FIG. 7 achieves the same result with a receivebeamformer 122 that demodulates the receive signals from the transducerarray to emphasize the harmonic 2f₀ (schematically indicated by thefilter 124). As described above, echo information from a single transmitevent is preferably applied both to the imaging and the correction datapaths, such that calculation of aberration correction values proceedsconcurrently with image formation.

Though the filters 112, 114 are shown as harmonic filters in FIG. 6,they do not have to be precisely the same filters. In fact, in oneparticularly preferred embodiment the filter 114 for the image processorpasses as wide a bandwidth as possible about the second harmonic whilestill rejecting the fundamental frequency band, and the filter 112passes a narrower pass band (possibly as much as a factor of two or morenarrower than that used in the filter 114). This narrower bandwidth maybe beneficial in the conversion from phase delay to time delay in theevent the aberration correction value estimator 26 uses a narrow bandcorrelation technique (i.e. measuring phase difference between adjacentelements or subarrays) to estimate time delays.

FIGS. 8 through 12 relate to other embodiments of the receive signalprocessor of FIG. 1 that are particularly well adapted to allow readyadjustment of the frequency bandpass used for the correction path andthe imaging path. As shown in FIG. 8, the receive signal processor 130includes a single beam receive beamformer 132 that provides at itsoutput a wideband receive signal that includes both the fundamental andthe harmonic components. This output signal of the receive beamformer132 is applied in parallel to m different filters 134 a-134 m. Each ofthe filters 134 a-134 m is adjustable, and the pass band of therespective filters can be controlled by respective control inputs. Theoutputs of the various filters 134 a-134 m are applied to separateaberration correction value estimators 138 a-138 m. Each aberrationcorrection value estimator 138 a-138 m generates a respective set ofaberration correction values based on the respective filtered inputsignal. Selected ones of these aberration correction values are furtherprocessed, as controlled by the switches 140 a-140 m. Though not shownin FIG. 8, a partial summation as described above can be provided in thecorrection path.

As shown in FIG. 8, the imaging path includes a summer 142 that providessummed, beamformed signals as inputs to a plurality of imaging filters144 a-144 n. Each of the imaging filters 144 a-144 n is separatelyadjustable as to pass band by a respective control input, and theoutputs of selected ones of the filters 144 a-144 n are passed to theimage processor 24, gated by switches 146 a-146 n. This arrangementallows great flexibility in choosing one or more signals of respectivefrequency bands for aberration correction value estimation and forimaging. For example, the image processor 24 can combine the signalsfrom multiple frequency bands in any desired manner. If desired, thereceive beamformer 132 can be of the type that provides multiple receivebeams, demodulated at respective center frequencies.

The receive signal processor 150 of FIG. 9 supplies the harmoniccomponent of the ultrasound echo information to the image processor, asfiltered by the pass band filter 152. The correction path in the receivesignal processor 150 includes two separate pass band filters 154, 156connected in parallel. In this embodiment the filter 154 is centered atthe fundamental pass band, and the filter 156 is centered at theharmonic pass band. The outputs of these filters 154, 156 are applied toseparate respective aberration correction value estimators 158, 160. Theembodiment of FIG. 9 supports imaging at the harmonic and aberrationcorrection using a combination of harmonic and fundamental components ofthe echo information. It is anticipated that it may be particularlydesirable to combine the fundamental frequency component (which providesexcellent penetration and reduced signal to noise problems) with theharmonic component for improved accuracy because of the smaller transmitbeam profile. For example, an average, a weighted average or athresholded average of the two components may be used.

FIG. 10 shows another receive signal processor 170 that includes a dualbeam receive beamformer 172. The two beams are demodulated with centerfrequencies at the fundamental and second harmonic, respectively.Ultrasound echo information from either of the two receive beamsgenerated by the receive beamformer 172 can be routed to the correctionpath via the switch 171 and the filter 174. Similarly, ultrasonic echoinformation in either of the two receive beams can be routed via theswitch 175, the summer 176, the filter 178 and the switch 179 to theimage processor. By properly controlling the switches of FIG. 10 eitherone of the two receive beams can be used as a source of ultrasound echoinformation for either of the two signal paths.

FIG. 11 shows another receive signal processor 190 that includes asingle beam receive beamformer 192 that is demodulated in a broadbandmariner such that both fundamental and harmonic components appear at theoutput of the receive beamformer. The correction signal path includes anadjustable filter 194 and a switch 195. The image signal path includesan adjustable filter 196 and a switch 198. The receive signal processor190 of FIG. 11 can be used in a time multiplexed fashion. By properlycontrolling the adjustable filters and the switches the ultrasound echoinformation can be routed differently for different firings of thetransducer.

The processors of FIGS. 10 and 11 may be operated in a sequential modein which the signals for aberration correction estimation are firstfiltered with one filter configuration and the resulting aberrationcorrection values stored in correction value memory, and then aberrationcorrection values are created in a second transducer firing from asignal filtered with a different filter configuration, or an all passfilter (i.e. no filter). Once the second aberration correction value isobtained, these two estimates (both stored in the correction valuememory) are passed to the adaptive focus processor 28 of FIG. 1, whichanalyzes the results of the estimates of the aberration correctionvalue. The advantage of using more than one estimate of aberrationcorrection value is that a more stable and reliable result may beobtained. In one version, only aberration correction values derived fromthe fundamental component of the received signal are analyzed.

As another example, the transducer may be fired at a fundamentalfrequency band and then ultrasound echo information may be selectivelyreceived from this first firing at a harmonic frequency band and usedfor image processing. Then a second ultrasound firing can be performedat the fundamental frequency band, and received ultrasonic informationfrom the second firing in the fundamental frequency band can be used todetermine aberration correction values. Alternatively the harmonicfrequency band from the second firing or both the harmonic andfundamental frequency bands from the second firing can be used todetermine aberration correction values. It is not essential in all casesthat fundamental and harmonic frequency bands be used, and in analternative embodiment a lower frequency band can be transmitted in thefirst and second firings. A higher frequency band can be used for atleast a portion of the image and either the higher, the lower, or bothfrequency bands can be used in the correction path for the calculationof aberration correction values.

FIG. 12 shows another receive signal processor 210 that is designed tooperate with consecutive firings of the transducer at differentfrequency bands. The receive signal processor 210 includes a receivebeamformer 212 that supplies receive signals to a correction path via anadjustable filter 214 and to an imaging path via an adjustable filter216. Switches 218, 220 are provided in each path. The receive signalprocessor 210 can be used in a mode in which the transducer array firsttransmits an ultrasound signal centered at the fundamental pass band attime T₀, and then later transmits ultrasound information at time T₁ atthe harmonic pass band 2f₀. Thus the output signal of the receivebeamformer 212 at time T₀ includes both fundamental and harmoniccomponents, while the output signal associated with the firing at timeT₁ includes harmonic components without fundamental components.Depending upon the adjustment of the filters 214, 216 and the control ofthe switches 218, 220 many alternatives are possible. For example,information acquired in response to either of the first or the secondfirings at times T₀ and T₁, respectively, can be filtered to isolate theharmonic component which is applied to the image processor. The filter214 and the switch 218 can be controlled to pass the harmonic componentfrom the signal associated with the firing at time T₁ to the aberrationcorrection value estimator, or the fundamental component from the signalassociated with the firing at time T₀ to the aberration correction valueestimator.

System Optimization Processor

As discussed above, the receive signal processor can be configured inmany ways depending upon the application. For this reason, it will oftenbe preferable to provide a system for optimizing operation of thereceive signal processor to select the most appropriate mode ofoperation for the imaging application of interest.

One approach is to allow the user to select one of a number ofpre-programmed configurations for the receive signal processor. Eachconfiguration defines the number of beams to be formed by the receivebeamformer, the filters to be used (or the filter parameters foradjustable filters), and the routing of signals to the imaging andcorrection paths, i.e. which switches should be closed. For example, thearchitecture of FIG. 14 can be programmed to operate in any of fourdifferent modes, as indicated by the four rows of Table 1. With thisembodiment the user selects one of the four settings of Table 1 as amatter of preference when imaging the patient.

TABLE 1 Number of Beam number receive selected for Selection Filter 54Filter 58 Filter 60 Filter 64 beams correction path 1 2f₀ WB 2f₀ MB  f₀WB  f₀ NB 2 2 2  f₀ WB  f₀ MB 2f₀ WB 2f₀ NB 2 2 3 2f₀ WB 2f₀ MB n/a 2f₀NB 1 1 4  f₀ WB  f₀ MB n/a  f₀ NB 1 1 WB = wider band (e.g., 2 times BWof MB) MB = medium band (e.g., 2 times BW of NB) NB = narrower band(e.g., ¼ to ½ of f₀)

A second approach to system optimization is to use pre-set parametersfor the receive signal processor that are selected automatically inresponse to the mode of operation of the imaging system. For example,lookup tables can be provided storing the parameters to be used for thereceive signal processor under various scanning conditions. Differenttables or entries in a single table can be accessed and used to programthe receive signal processor depending upon the following parameters:(a) the mode of imaging (e.g. B-mode, M-mode, color Doppler mode,Doppler tissue imaging mode, etc.); (b) the transducer being used (e.g.a transesophageal transducer may best be used with the harmoniccomponent applied to both the imaging and correction paths); (c) thedepths within the image from which the echoes are being returned forboth the image and the region at which aberration correction values arebeing calculated. In this embodiment, system parameters for the receivesignal processor are automatically selected based on empirical studiespre-programmed for the given scanning conditions.

For example, aliasing and signal to noise ratios can be determined byroutine experimentation, and the results of these experiments can beused to select the receive signal processor programming parametersstored in the lookup tables. Table 2 provides a rough estimate of signallevels associated with various types of signals supplied by the receivesignal processor discussed above.

TABLE 2 Imaging/Correction Mode Relative SNR Level B-mode imaging at  0dB fundamental subarray correction data −15 dB at fundamental B-modeimaging at −25 dB harmonic F-mode imaging at −40 dB fundamental subarraycorrection data −40 dB at harmonic F-mode imaging at −65 dB harmonic

A third approach to optimizing a programmable receive signal processoris an automatic, adaptive system as shown in FIG. 13. In FIG. 13 thereceive signal processor 230 includes a single beam receive beamformer232, two parallel filters 234, 236 in the correction signal path and twoparallel filters 238, 240 in the imaging signal path. Also included is asystem optimization processor 242 that controls selection of one of thetwo filters 234, 236 and one of the two filters 238, 240. In thisembodiment the output of the receive beamformer 232 is a broadbandsignal that includes both harmonic and fundamental components of thereceive signal. By way of example, the filters 234, 238 may selectivelypass the fundamental frequency component and the filters 236, 240 mayselectively pass the harmonic frequency component. The systemoptimization processor 242 also receives two estimates of quality. Thefirst estimate of quality is provided on line 244 from the aberrationcorrection value estimator. This signal is a real time measure of thequality of the aberration correction values. The second input signal tothe system optimization processor 242 is an image quality signal appliedvia line 246 from the image processor. The signal on line 246 is a realtime measure of the quality of the image. By way of example, the qualityestimates may simply be measures of the signal levels in the image andsignal levels applied to the aberration correction value estimator. Thesystem optimization processor uses the quality estimates on lines 244and 246 to vary filter selection. The estimates of quality may beevaluated at one depth such as the gating depth for the aberrationcorrection system or preferably at various depths throughout the image.

FIG. 14 shows a block diagram of another receive signal processor 250 inwhich the system optimization processor 252 controls beam selection froma dual beam receive beamformer 254, filter selection for the individualbeams within the beamformer 254, and control settings for the filters256, 258 in the correction and imaging paths, respectively. In theembodiment of FIG. 14, the system optimization processor 252 optimizesthe filter settings, the number of beams, and the selection of beamsbased on real time quality measurements. In many cases, the systemoptimization processor 252, 242 will supply information indicative ofthe frequency component of the information applied to the imageprocessor to facilitate effective image processing such as scanconversion.

Alternate Filtering Techniques

In an alternate embodiment, the following beamforming technique (knownhereafter as alternate channel phasing) may be used as an alternative ora supplemental method for suppressing or filtering the fundamentalsignal energy. In the receive beamformer 82 of FIG. 4, the focusingdelays used to calculate the second harmonic receive signals passed tothe aberration value estimator are modified, so that every other channelhas an additional delay equal to one cycle of the harmonic centerfrequency. In addition, the partial summers 92 are configured to sumsignals received by an even number of channels. This delay represents a2*pi radian phase shift at the harmonic frequency, so it has little orno impact on the harmonic signal. However, it represents a pi radianphase shift at the fundamental frequency. Since half of the summedchannels experience no phase shift at the fundamental, while the otherhalf of the summed channels experience a pi radian phase shift, thefundamental signal at the sum is effectively canceled. This may be ofadvantage, since it can be expensive to design and produce filters 86for each received channel or filters 94 for each subarray signal whichadequately remove the fundamental component of the received signal. Thisbeam formation technique may either reduce the need to performindividual filtering, or increase the overall rejection of thefundamental (i.e., the degree to which the received fundamental energyis removed from the signal).

Numerous variations on this alternate channel scheme are possible. Forexample, the partial summers 92 of FIG. 4 may sum an odd number ofchannels, such as three. In this case, the fundamental would be onlyimperfectly canceled (for example, summing two channels with a pi phaseshift and one channel with no phase shift only imperfectly cancels thefundamental). The degree of fundamental rejection may be improved byweighting the individual channel signals before summing. In thethree-channel sum case, the center channel signal may be weighted twiceas much as the outer channel signals.

Many other permutations are possible. Delays other than T₂, where T₂ isthe period of the harmonic frequency, may be used, optionally inconjunction with weighting of the individual channel signals beforesumming. For example, in the three channel sum case, if a delay of5/6*T2 is used on the two outer channels, and those same outer channelsare weighted by a factor of 0.5774 before summing (where the centerchannel is weighted with a factor of 1), then the fundamental signalcomponents of the summed signal will cancel, while the harmoniccomponents are only slightly degraded. A technique such as this may beof use if the time delays that can be applied to the various channelsare constrained.

This technique can be used to good effect in many of the otherembodiments described herein. For example, in the architecture of FIG.6, wherein both imaging and aberration estimation are performed at theharmonic frequency, alternate channel phasing may be used to improve thefundamental rejection of both the correction measurement signal and theimaging signal. In the architecture of FIG. 2, the technique can be usedto improve fundamental rejection of the imaging signal (at the harmonicfrequency) only, while the correction signal uses unperturbed focusingdelays.

Less preferably, in the architectures of FIGS. 2 and 3, the techniquecan be used for both the imaging (harmonic frequency) and correction(fundamental frequency) paths. This may function adequately if: (1)aberration is estimated from individual element signals, or on subarrayssummed from an odd number of elements, so that the cancellation isimperfect (in which case, the estimated aberration values should becorrected for the effects of the additional delays); or (2) the receivedfundamental signals are so much larger than the received harmonicsignals that even after cancellation of the fundamental by the alternatechannel phasing technique, there remains sufficient energy at thefundamental frequency. (Typically, the fundamental signal is expected tobe 25-30 dB higher than the harmonic, so this case may well occur.)

Additive Inverse Embodiments

In alternate embodiments of the signal processor, the additive inversetechnique is used in place of or in addition to filtering to improve theseparation between linear components of a signal (components derivedfrom linear propagation and scattering) and non-linear components(components derived from non-linear propagation or scattering such assecond harmonic signals). The additive inverse technique is discussed inChapman U.S. Pat. No. 5,632,277 and Hwang U.S. Pat. No. 5,706,819. Thetechnique is discussed in further detail and further variations of thetechnique are described in U.S. patent application Ser. No. 09/061,083,filed Apr. 15, 1998, now U.S. Pat. No. 5,902,243, filed concurrentlywith this application and hereby incorporated by reference. As explainedbelow, the additive inverse technique can be used to generate harmonicsubarray signals from which aberration corrections are derived, or fromwhich harmonic images are generated.

In the simplest embodiment of the additive inverse technique, twosuccessive transmit pulses are fired along a single ultrasound line. Thetwo pulses are identical, except that the second pulse is inverted withrespect to the first pulse. For a modulated sinusoidal pulse, this isequivalent to changing the phase of the sinusoid by pi radians. Forexample, the two pulses may be represented as follows:

P 1=e(t)*cos(2*pi*f 1*t),

P 2=e(t)*cos(2*pi*f 1*t+pi),

where e(t) represents the pulse envelope and f1 represents the transmitcenter frequency. From the two transmit firings, two receive waveformsRa(t) and Rb(t) are acquired. The receive waveforms are acquired as RF,IF, or complex baseband data, and are considered here prior to anydetection operation that would remove phase information and convert thereceive signal to an intensity. In this specification such pre-detectionsignals will be referred to as “analytic” waveforms.

In general, the analytic waveforms of the receive signals associatedwith the two transmit events comprise fundamental and harmoniccomponents. The two signals are in one sense quite similar. However, thefundamental and odd harmonic components of the two signals will beinverted as with the transmit pulses, while the even harmonic componentswill not be inverted:

Ra(t)=R 1(t)+R 2(t)+R 3(t) . . . ,

Rb(t)=−R 1(t)+R 2(t)−R 3(t) . . . ,

where R1(t), R2(t), and R3(t) are the fundamental, second harmonic, andthird harmonic components, respectively, of the receive signals Ra(t),Rb(t). When the two receive signals Ra(t) and Rb(t) are added, thefundamental and odd harmonic components of the receive signals cancel,leaving primarily components arising from second harmonic propagationand scattering. When the two receive signals Ra(t), Rb(t) aresubtracted, the even harmonic components cancel, and the resultingdifference contains primarily components arising from linear propagationand scattering. Many variations of the additive inverse technique can beapplied in conjunction with the adaptation techniques disclosed in thepresent specification, including those described in the above-identifiedrelated application. These variations include, but are not limited to,(1) the combination of more than two pulses to form a combined signal,the use of phase differences other than pi radians, the use of transmitpulses comprising both inverted and non-inverted components, and the useof unipolar transmit pulses, all as described in above-referenced U.S.patent application Ser. No. 09/089,467, filed Jun. 2, 1998, nowabandoned, and (2) the use of spatially distinct transmit or receivebeams, as described in U.S. patent applications Ser. Nos. 08/993,395 and08/993,533.

FIG. 18 shows one receive signal processor 300 that combines additiveinverse techniques with adaptive focus correction techniques. Theultrasound imaging system that includes the signal processor 300generates two transmit pulses of alternate polarity. The receivebeamformer 302 generates as outputs analytic signals via optionalsubarray summer 302. The transducer element signals or the subarraysignals from the first transmit firing are stored in an analytic linebuffer 304. The transducer element or subarray signals from the secondtransmit firing are then added to the stored signals from the firsttransmit firing in a summer 306, using the same summing polarity forboth signals to constructively reinforce the second harmonic signals orcomponents and to cancel the fundamental components. The resultingcombined or summed signal is then applied to the aberration correctionpath that includes the aberration correction value estimator 26,optionally via filter 308. In this embodiment the filter 308 is apassband filter arranged to pass second harmonic components whileblocking fundamental components. The filtration requirements on thefilter 308 are less severe than they would be if the additive inversetechnique were not used. The filter 308 may be omitted or may be broaderband or less deep than would otherwise be required, because asignificant part of the linear components of the subarray or transducerelement signals are cancelled by the additive inverse summing operation.This may be of particular advantage, as any filter 308 is repeatedacross all elements or all subarrays of the system. It should be notedthat the analytic line buffer 304 need not store the entire receivesignal for each element or subarray. Only a subset of samples (coveringa limited set of ranges at which aberration correction values are to beestimated) needs to be stored and then processed by the aberrationcorrection value estimator 26. In addition, the receive signals can besubsampled prior to storage.

On the imaging path of the signal processor 300, receive signals fromthe beamformer 302 are routed to beam summer 310 and are then filteredby an optional filter 312. The receive signals from the first transmitfiring are stored in analytic line buffer 314. The stored signals frombuffer 314 are then summed with the receive signals from the secondtransmit firing with opposite summing polarity in summer 316. Theresulting difference signal is then stored, scan converted, anddisplayed by the image processor 24. Because the summer 316 operateswith opposite summing polarities to emphasize the fundamental andsuppress the second harmonic component, the filter 312 is a low-pass orpassband filter designed to pass the fundamental components whileblocking second harmonic components.

In general, the buffer 314 and summer 316 may be incorporated into theintensity processing block 36 or the motion estimator block 38 of FIG.1. The buffer 314 and the summer 316 may also be incorporated prior tothe filter 312 or prior to the beam summer 310. This may, however,increase the required size of the line buffer 314. The buffer 314 andthe summer 316 may be omitted in this embodiment, as the receive signalsprior to the summer 316 usually are dominated by the fundamental(linear) components of the receive signal. However, by combining the twosignals the signal to noise ratio of the combined signal is increasedand is therefore beneficial in some applications.

FIG. 19 provides a flowchart of the operation of an ultrasonic imagingsystem including the receive signal processor 300 of FIG. 18. In step330 multiple ultrasound pulses are fired into a target with fundamentalcomponents that differ in phase by about pi radians. In step 332 receivesignals are formed from echoes of the ultrasound pulses of step 330, andin step 334 first and second receive signals are summed with common oropposite summing polarities to form a combined signal. In step 336 thecombined signal and/or selected receive signals are applied to the imageprocessor, to the aberration correction value estimator, or to both.

Though FIG. 18 shows one choice of signal paths and frequency bands,many others are possible. Either the fundamental or the harmonicfrequency band can be applied to either the aberration correction pathor the imaging signal path. Also or alternatively, a single receivesignal prior to summation may be applied to either signal path. Eitheror both of the filters 308, 312 may be deleted if desired, and if usedthey may be placed either before or after the summer 306, 316. The sameor different frequency bands can be applied to the imaging andaberration correction signal paths.

FIG. 20 shows a block diagram of another receive signal processor 350that provides additional flexibility. The signal processor 350 isintended for use in the system of FIG. 1 described above. In theprocessor 350, receive signals from a receive beamformer (includingreceive signals from first and second transmit events that differ inphase angle by about pi radians) are stored in a memory 352.Time-aligned pairs of samples (one from the first pulse and one from thesecond pulse) are then retrieved from the memory 352 and applied to asummer 354. The summer 354 is controllable by a function control input355 to sum the two inputs with either common summing polarity oropposite summing polarity as desired. The combined signal generated bythe summer 354 is then optionally passed through a programmable filter356 and via a switch 358 to either the image processor 24 or theaberration correction value estimator 26. The filter 356 is not requiredin all embodiments. If included as a programmable filter, it can beadjusted to selectively pass the second harmonic component or thefundamental component, depending upon the state of the control signal355. If desired, the switch 358 can be deleted, and each signal path canbe provided with a respective summer and optional filter.

FIG. 21 provides a flow chart of the operation of an ultrasonic imagingsystem including the receive signal processor 350 of FIG. 20. In step360 first and second ultrasound pulses are fired into a target withfundamental components that differ in phase by pi radians. In step 362first and second receive signals are formed from echoes of the first andsecond pulses. These receive signals are stored in the memory of 352 ofFIG. 20 in step 364. The stored signals from memory 352 are then summedwith common polarity in summer 354 (step 366) and with oppositepolarities (step 368). Then at least one of the summed or combinedsignals is applied to an image processor and/or an aberration correctionvalue estimator (step 370) via the filter 356 and the switch 358 of FIG.20.

By storing both receive signals in the memory 352, the embodiment ofFIG. 20 allows both summed (non-linear) and substracted (linear)combined signals to be generated from a single set of received data.This flexibility can be used to good effect, for example by allowing asingle aberration correction value estimator to estimate corrections atboth a harmonic and a fundamental frequency from a single pair offirings. Alternately, a single pair of firings can be used to supply theharmonic signals, the fundamental signals, or both to the imageprocessor 24 and the harmonic signals, the fundamental signals, or bothto the aberration correction value estimator 26.

The additive inverse techniques described above can be applied to eitherthe aberration correction value path or the imaging path or both in anyof the specific embodiments disclosed in the preceding FIGS. 1-17 Wherefilters are included they can be made programmable or not, depending onthe application, and they can be positioned before or after the varioussummation steps. The fundamental components can be applied to either theimaging path, the aberration correction path, or both. Similarly theharmonic components can be applied to either the imaging path, theaberration correction path, or both.

As explained above, these embodiments are not limited to the simplestversions of the additive inverse technique as described above, but canbe also used with more complicated versions, as explained in co-pendingU.S. patent application Ser. No. 09/061,083.

As used herein the term “cancel” is intended broadly to encompasspartial cancellation. The term “applying” is intended broadly toencompass both direct application (where a signal is applied to adownstream processor without modification) and indirect application(where a signal is modified before it is applied to a downstreamprocessor).

Definitions

The terms discussed below are intended to be given the followingmeanings, both in this specification and in the following claims.

Fundamental components or information correspond to signals (whethermodulated or demodulated) associated with or responsive to signals in afundamental frequency band when ultrasonically modulated as an acousticsignal. Similarly, harmonic components or information correspond tosignals (whether modulated or demodulated) associated with or responsiveto signals in a harmonic frequency band when ultrasonically modulated asan acoustic signal. For example, a harmonic component may be modulatedat a harmonic ultrasonic frequency such as 5 MHz, or it may be a DCsignal that is responsive to the portion of a signal that was modulatedat the harmonic frequency prior to demodulation. Of course, bothfundamental and harmonic components will generally include measurableamounts of both. For example, FIG. 16 shows a frequency spectrum for anultrasound signal as applied to any of the receive signal processorsdescribed above. Most of the ultrasonic energy is near the fundamentalfrequency f₀, and the harmonic peak at 2f₀ has a peak amplitude that isreduced by 10 to 40 dB (e.g. 25 dB) from the peak amplitude at f₀. Theharmonic component generated by the receive signal processor asdescribed above may, for example, have a frequency spectrum as shown inFIG. 17, where the peak at f₀ is reduced with respect to the peak at 2f₀by about 3 to 40 dB (e.g. 15 dB).

Fundamental filters or pass bands selectively pass fundamentalcomponents while suppressing harmonic components, while harmonic filtersor pass bands selectively pass harmonic components while suppressingfundamental components. For example, a harmonic filter may be a passband filter centered at 5 MHz for use with a modulated signal or a lowpass filter for use with a demodulated signal.

The term “pass band” is intended broadly to include low pass and highpass as well as conventional pass bands, and the frequency pass band ofa signal path refers to the frequency of signals modulated at ultrasoundfrequencies, prior to any demodulation, regardless of whether signalsleave the path in modulated or demodulated form.

“Responsive to” or “in response to” is intended broadly to encompassresponsiveness, whether direct or indirect. Thus, a circuit isresponsive to a signal, whether or not the signal is subjected tointermediate processing prior to being applied to the circuit.

The term “portion of an image” is intended broadly to encompass anysignal that contributes to image formation, whether or not other signalsare used at other times, or for other parts of the image, or incombination with the first signal.

The term “filter” is intended to encompass any means for suppressingsignals outside a selected frequency band. Examples of filters includepass band filters, low pass filters, demodulators, and alternate channelphasing circuits, as described above.

The terms “selectively receiving” or “selectively transmitting” meanthat substantially a majority of the energy that is processed, received,or transmitted is responsive to or corresponds to energy within aselected band of frequencies in the acoustic domain. For example,selectively receiving a harmonic component means passing the harmoniccomponent while suppressing the fundamental component. Any of thefiltering techniques described above can be used, either alone or invarious combinations.

The term “harmonic” includes integer harmonics (e.g. second, third) aswell as fractional harmonics and subharmonics, unless otherwiseindicated.

The term “aberration correction value” is intended broadly to includefocusing (phase and/or delay) and/or apodization corrections, unlessotherwise indicated.

It should be understood that the foregoing detailed description hasdescribed only a few of the many forms that the present invention cantake. It is therefore intended that only the following claims, includingall equivalents, be regarded as a definition of the invention.

What is claimed is:
 1. An ultrasonic imaging method comprising thefollowing steps: (a) transmitting an ultrasound signal; (b) receiving awide band ultrasound echo signal; (c) filtering the echo signal to forma first low frequency component and a second high frequency component;(d) forming at least part of an image at least in part in response tothe second high frequency component; (e) determining aberrationcorrection values at least in part in response to the first lowfrequency component.
 2. The method of claim 1 further comprising thestep of maintaining the target free of additional contrast agentthroughout an ultrasound imaging examination session that includes steps(a)-(e).
 3. An ultrasonic signal processing method comprising the steps:(a) transmitting ultrasonic energy into a target in first and secondtransmit events, said ultrasonic energy comprising at least afundamental component that differs in phase in the first and secondtransmit events by a selected phase angle; (b) forming first and secondreceive signals associated with the first and second transmit events,respectively; (c) combining the first and second receive signals to forma combined signal; and (d) applying the combined signal to an aberrationcorrection value estimator; wherein step (c) comprises the step ofsumming the first and second receive signals with opposite summingpolarities to cancel components of the combined signal in a frequencyband including a second harmonic of the fundamental component.